CN114006613A - Low-Power Interface Circuits for MEMS Sensors - Google Patents
Low-Power Interface Circuits for MEMS Sensors Download PDFInfo
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03K—PULSE TECHNIQUE
- H03K19/00—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits
- H03K19/0175—Coupling arrangements; Interface arrangements
- H03K19/017509—Interface arrangements
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- G—PHYSICS
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- G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
- G01D18/00—Testing or calibrating apparatus or arrangements provided for in groups G01D1/00 - G01D15/00
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- G—PHYSICS
- G01—MEASURING; TESTING
- G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
- G01D5/00—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable
- G01D5/12—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means
- G01D5/14—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage
- G01D5/20—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying inductance, e.g. by a movable armature
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- G01D5/00—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable
- G01D5/12—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means
- G01D5/14—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage
- G01D5/24—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying capacitance
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Abstract
本发明公开了一种用于MEMS传感器的低功耗接口电路，低功耗接口电路包括依次电连接的压控振荡单元、波形转换单元和频率电压转换单元，MEMS传感器用于采集外部设备的物理参量，并将物理参量的变化值转换为电容变化值ΔC或电感变化值ΔL，压控振荡单元用于将电容变化值ΔC或电感变化值ΔL转换为频率变化值ΔF，波形转换单元用于将频率变化值ΔF转换为时间周期变化值ΔT，频率电压转换单元用于依据时间周期变化值ΔT，呈时间周期变化地输出测量电压VO；本发明的电路结构简单，对工作环境要求较低，且能够对测量的灵敏度、动态范围和标称点进行灵活调节，有效在单一低功耗接口电路上满足对MEMS传感器为电容式传感器或电感式传感器的测量。
The invention discloses a low power consumption interface circuit for a MEMS sensor. The low power consumption interface circuit comprises a voltage-controlled oscillation unit, a waveform conversion unit and a frequency-to-voltage conversion unit that are electrically connected in sequence, and the MEMS sensor is used for collecting physical data of external equipment. The change value of the physical parameter is converted into the capacitance change value ΔC or the inductance change value ΔL, the voltage-controlled oscillation unit is used to convert the capacitance change value ΔC or the inductance change value ΔL into the frequency change value ΔF, and the waveform conversion unit is used to convert the The frequency change value ΔF is converted into a time period change value ΔT, and the frequency-voltage conversion unit is used to output the measurement voltage VO in a time period change according to the time period change value ΔT; the circuit structure of the present invention is simple, the requirements for the working environment are low, and The sensitivity, dynamic range and nominal point of the measurement can be flexibly adjusted, effectively satisfying the measurement of the MEMS sensor being a capacitive sensor or an inductive sensor on a single low-power interface circuit.
Description
Technical Field
The invention relates to the technical field of micro electro mechanical systems, in particular to a low-power-consumption interface circuit for an MEMS sensor.
Background
MEMS sensors have been widely found in various everyday electrical devices, including capacitive sensors and inductive sensors. The capacitive sensor is mainly applied to measurement of displacement, angle, vibration, speed, pressure, component analysis, medium characteristics and the like. Capacitive sensors basically use various types of capacitors as sensing elements to convert a physical or mechanical quantity to be measured into a change in capacitance, which is equivalent to a variable capacitor. A disadvantage of capacitive sensors is the need to ensure that the environment to be measured is free of contaminants such as dust, oil and water, as these factors can alter the dielectric constant and thus the measurement results. In addition, since the distance relationship between the capacitance of the capacitive sensor and the electrodes is nonlinear, the parasitic capacitance is large, the output impedance is high, and it is necessary to compensate the output for various environmental factors such as voltage, temperature, humidity, etc. depending on the compensation circuit.
The inductive sensor is mainly used for positioning metal objects in a short distance and is widely applied to industries such as automobile manufacturing, robots and the like. The inductance type sensor is a device for realizing measurement by utilizing the change of coil self-inductance or mutual inductance, has simple structure, no movable electric contact, long service life, high sensitivity and resolution, strong output signal and better linearity and repeatability, can realize the remote transmission, recording, display and control of information, and can measure parameters such as displacement, vibration, pressure flow, specific gravity and the like.
The interface circuit of the capacitive or inductive sensor mainly has two types: amplifier-based and modulation-based configurations. Amplifier-based circuits can be generalized into three categories: an ac bridge, a transimpedance and a switched capacitor. These circuits provide an amplified voltage change proportional to the sensor capacitance change. However, various noise sources such as: flicker noise, thermal noise, substrate noise coupling, etc., and parasitic capacitance, have a certain effect on the readout range and resolution of the sensor.
The sensor interface circuit with the modulation circuit can effectively reduce the noise level of noise and direct current offset in the circuit. The modulation circuit has a variety of: sigma-delta converters, successive approximation analog-to-digital converters, chopper modulation, and the like. Sigma-delta converters require higher speed analog circuitry because the sampling rate is much higher than the effective bandwidth, and in addition, the voltage dependent nonlinear effects of the capacitors also result in lower signal-to-noise ratios; successive approximation analog-to-digital converters use a search algorithm to compare the sensor analog signal with the analog signal generated sequentially from the digital-to-analog converter, however this design requires a higher number of bits of analog-to-digital converter; chopper modulation is an amplitude modulation technique which takes square waves as carrier signals, and is used for shifting a sensor signal to a higher frequency to suppress noise, then demodulating an amplified modulation signal and recovering an original signal through a low-pass filter.
Disclosure of Invention
The invention aims to provide a low-power-consumption interface circuit for an MEMS sensor, which has a simple circuit structure and lower requirements on a working environment, can flexibly adjust the sensitivity, the dynamic range and the nominal point of measurement, and effectively meets the requirement that the MEMS sensor is a capacitive sensor or an inductive sensor on a single low-power-consumption interface circuit.
In order to achieve the purpose, the invention discloses a low-power-consumption interface circuit for an MEMS sensor, which comprises a voltage-controlled oscillation unit, a waveform conversion unit and a frequency-voltage conversion unit which are sequentially and electrically connected, wherein the MEMS sensor is used for acquiring physical parameters of external equipment and converting the change value of the physical parameters into a capacitance change value delta C or an inductance change value delta L, the voltage-controlled oscillation unit is used for converting the capacitance change value delta C or the inductance change value delta L into a frequency change value delta F, the waveform conversion unit is used for converting the frequency change value delta F into a time period change value delta T, and the frequency-voltage conversion unit is used for outputting a measurement voltage VO in a time period change manner according to the time period change value delta T.
Compared with the prior art, the voltage-controlled oscillation unit is used for converting a capacitance change value delta C or an inductance change value delta L into a frequency change value delta F, the waveform conversion unit is used for converting the frequency change value delta F into a time period change value delta T, the frequency-voltage conversion unit is used for outputting a measurement voltage VO in a time period changing manner according to the time period change value delta T, the circuit structure is simple, the requirement on the working environment is low, the measurement sensitivity, the dynamic range and the nominal point can be flexibly adjusted, and the requirement that an MEMS sensor is a capacitive sensor or an inductive sensor on a single low-power-consumption interface circuit is effectively met.
Preferably, the MEMS sensor is a capacitive sensor or an inductive sensor, and when the MEMS sensor is a capacitive sensor, the MEMS sensor is equivalent to a sensing capacitor CS, and when the MEMS sensor is an inductive sensor, the MEMS sensor is equivalent to a sensing inductor LS.
Preferably, the voltage-controlled oscillation unit is a differential cross-coupled voltage-controlled oscillator.
Preferably, the voltage-controlled oscillation unit comprises a first field-effect transistor M1, a second field-effect transistor M2, a third field-effect transistor M3, a fourth field-effect transistor M4, a varactor diode circuit and a resonant inductor L0, wherein the first field-effect transistor M1, the second field-effect transistor M2, the third field-effect transistor M3 and the fourth field-effect transistor M4 jointly form a differential cross-coupled voltage-controlled oscillation circuit, the varactor diode circuit is connected in parallel with the differential cross-coupled voltage-controlled oscillation circuit, and when the MEMS sensor is a capacitive sensor, the sensing capacitor CS is connected in parallel with the differential cross-coupled voltage-controlled oscillation circuit; when the MEMS sensor is an inductive sensor, the sensing inductor LS is connected in series with the resonant inductor L0, and then connected in parallel with the differential cross-coupled voltage-controlled oscillation circuit.
Specifically, the varactor circuit includes a fifth field effect transistor M5 and a sixth field effect transistor M6.
Preferably, the waveform converter is a sine-square wave converter.
Specifically, the waveform converter includes a cross-coupled inverter for converting a sine wave generated by the voltage controlled oscillation unit into a square wave, and a bias circuit for supplying a bias voltage to the voltage controlled oscillation unit.
Preferably, the voltage-controlled oscillation unit and the waveform conversion unit have the same output frequency.
Preferably, the frequency-voltage conversion unit includes a logic control module, a charge pump module and a feedback module, and the logic control module receives the time period variation value Δ T and converts the time period variation value Δ T into a pulse variation value Δ P; the charge pump module receives the pulse change value delta P and outputs a measurement voltage VO according to the pulse change value delta P in a time period change manner; the feedback module collects a voltage change value delta VO of the measurement voltage VO and feeds the voltage change value delta VO back to the logic control module, and the logic control module adjusts the pulse change value delta P according to the voltage change value delta VO.
Specifically, the logic control module is provided with a first input end for inputting a control voltage VC, the charge pump module is provided with a second input end for inputting a charging current IC, and the control voltage VC and the charging current IC jointly adjust parameters of the charge pump module.
Drawings
FIG. 1 is a circuit block diagram of a low power interface circuit for a MEMS sensor of the present invention;
FIG. 2 is a circuit diagram of the voltage controlled oscillation unit of the present invention not connected to the MEMS sensor;
FIG. 3 is a circuit diagram of the voltage controlled oscillation unit of the present invention when the MEMS sensor is connected as a capacitive sensor;
FIG. 4 is a circuit diagram of the voltage controlled oscillation unit of the present invention when the MEMS sensor is connected as a capacitive sensor;
FIG. 5 is a circuit block diagram of a frequency-to-voltage conversion unit of the present invention;
fig. 6 is a circuit diagram of a charge pump module of the present invention.
FIG. 7 is a timing diagram of the logic control module of the present invention.
Detailed Description
In order to explain technical contents, structural features, and objects and effects of the present invention in detail, the following detailed description is given with reference to the accompanying drawings in conjunction with the embodiments.
Referring to fig. 1, the low power consumption interface circuit for the MEMS sensor 1 of the present embodiment is suitable for respectively measuring when the MEMS sensor 1 is a capacitive sensor or an inductive sensor, and the MEMS sensor 1 is a sensor capable of acquiring a physical parameter of an external device and converting a change value of the physical parameter into a capacitance change value Δ C or an inductance change value Δ L. It is understood that the MEMS sensor 1 can be equivalent to a variable capacitance or a variable inductance, the MEMS sensor 1 being equivalent to a sensing capacitance CS when the MEMS sensor 1 is a capacitive sensor, and the MEMS sensor 1 being equivalent to a sensing inductance LS when the MEMS sensor 1 is an inductive sensor.
Referring to fig. 1-4, the low power consumption interface circuit includes a voltage-controlled oscillation unit 10, a waveform conversion unit 20 and a frequency-voltage conversion unit 30 electrically connected in sequence, wherein the voltage-controlled oscillation unit 10 is configured to convert a capacitance variation value Δ C or an inductance variation value Δ L into a frequency variation value Δ F, the waveform conversion unit 20 is configured to convert the frequency variation value Δ F into a time period variation value Δ T, and the frequency-voltage conversion unit 30 is configured to output a measurement voltage VO according to the time period variation value Δ T in a time period varying manner.
Preferably, the voltage-controlled oscillation unit 10 is a differential cross-coupled voltage-controlled oscillator, and of course, the voltage-controlled oscillation unit 10 may also be another oscillator capable of meeting the operation requirement of the present embodiment, and the type of the voltage-controlled oscillation unit 10 is not limited herein.
Preferably, the voltage-controlled oscillation unit 10 includes a first fet M1, a second fet M2, a third fet M3, a fourth fet M4, a varactor circuit, and a resonant inductor L0, and the first fet M1, the second fet M2, the third fet M3, and the fourth fet M4 together form a differential cross-coupled voltage-controlled oscillation circuit and provide a negative transconductance required by the circuit.
Specifically, the varactor diode circuit includes a fifth field effect transistor M5 and a sixth field effect transistor M6, and the equivalent capacitance value of the varactor diode circuit is equivalent capacitance C0. The varactor circuit is connected in parallel with the differential cross-coupling voltage-controlled oscillation circuit, and the equivalent capacitor C0 and the resonant inductor L0 jointly form a resonant tank circuit of the differential cross-coupling voltage-controlled oscillation circuit. The value of the equivalent capacitor C0 is determined by the source voltages VB of the fifth fet M5 and the sixth fet M6. When the sensor capacitance CS is set to 0, the nominal frequency of the voltage-controlled oscillation unit 10 is F0. Fig. 2 shows a circuit diagram of the voltage-controlled oscillation unit 10 not accessing the MEMS sensor 1, and at this time, the voltage-controlled oscillation unit 10 reserves an a-B port when the MEMS sensor 1 to be accessed is a capacitive sensor and a C-D port when the MEMS sensor 1 to be accessed is an inductive sensor.
Fig. 3 shows a circuit diagram of the voltage controlled oscillation unit 10 when the MEMS sensor 1 to be measured is a capacitive sensor, where the sense capacitance CS is connected in parallel to the differential cross-coupled voltage controlled oscillation circuit into the a-B port and the sense inductance LS is connected in the air. At this time, the time period change value Δ T of the voltage controlled oscillation unit 10 is Δ CS/(2 × F0 × C0).
Fig. 4 shows a circuit diagram of the voltage-controlled oscillating unit 10 when the MEMS sensor 1 to be measured is an inductive sensor, the sensing inductor LS is connected in series with the resonant inductor L0 after being connected to the C-D port, and then connected in parallel with the differential cross-coupled voltage-controlled oscillating circuit, and the sensing capacitor CS is connected in the air. At this time, the time period change value Δ T of the voltage controlled oscillation unit 10 is Δ LS/(2 × F0 × C0).
Referring to fig. 1, the waveform converter is a sine-square wave converter. Specifically, the waveform converter includes a cross-coupled inverter for converting a sine wave generated by the voltage controlled oscillation unit 10 into a square wave, and a bias circuit for supplying a stable bias voltage to the voltage controlled oscillation unit 10. Preferably, the voltage controlled oscillation unit 10 and the waveform conversion unit 20 have the same output frequency to ensure the accuracy of the measurement.
Referring to fig. 1-7, the frequency-voltage conversion unit 30 includes a logic control module 31, a charge pump module 32 and a feedback module 33, wherein the logic control module 31 receives the time period variation value Δ T and converts the time period variation value Δ T into a pulse variation value Δ P; the charge pump module 32 receives the pulse variation value Δ P and outputs a measurement voltage VO that varies in a time period according to the pulse variation value Δ P; the feedback module 33 collects the voltage variation value Δ VO of the measured voltage VO and feeds the voltage variation value Δ VO back to the logic control module 31, and the logic control module 31 adjusts the pulse variation value Δ P according to the voltage variation value Δ VO.
Specifically, the logic control module 31 has a first input terminal for inputting the control voltage VC, and the charge pump module 32 has a second input terminal for inputting the charging current IC, and the control voltage VC and the charging current IC jointly adjust the parameters of the charge pump module 32.
The logic control module 31 generates a control pulse S1 and a control pulse S2, wherein the control pulse S1 and the control pulse S2 are non-overlapping control pulses. In a specific arrangement, the frequency of the control pulse S1 and the control pulse S2 is the same as the frequency of the output of the waveform converter, but the duty ratio is low, and the pulse width depends on the input frequency T, so as to control the charge pump circuit to provide a stable dc output voltage.
The logic control module 31 uses inverters and parallel capacitor inverters to generate fixed and variable amounts of time delay, respectively. Since the delay slope of the parallel capacitor inverter is small relative to the delay of the dc control voltage, its output pulse produces low jitter and noise. The pulse widths of the control pulse S1 and the control pulse S2 depend on the sense capacitance CS, the sense inductance LS, the nominal frequency of the voltage controlled oscillator, the variation of the varactor capacitance or fixed inductance of the resonant tank circuit, and the delay slope feedback delay of the shunt capacitance.
Since the control pulse S1 and the control pulse S2 are generated by a differential clock, the influence of the process, voltage, temperature, and other changes on the two delay circuits is similar, so that the pulse widths of the control pulse S1 and the control pulse S2 are equal. Fig. 7 shows a timing chart of the logic control module 31 of the present embodiment.
Referring to fig. 1-7, the charge pump circuit includes a first capacitor C1, a second capacitor C2, a seventh fet M7, an eighth fet M8, a ninth fet M9, a tenth fet M10, a first constant current source IC and a first constant current source ID, the seventh fet M7 and the eighth fet M8 are controlled by a control signal S1B and a control signal S2, respectively, and the ninth fet M9 and the tenth fet M10 function as transfer transistors in the charge pump circuit and are controlled by a control signal S1 and a control signal S1X, respectively.
In the first half period of T, the control signal S1 and the control signal S1X turn on the seventh fet M7, the ninth fet M9 and the tenth fet M10, the first capacitor C1 and the second capacitor C2 instantaneously redistribute the stored charges, and the first capacitor C1 and the second capacitor C2 are charged by the first constant current source IC during the turn-on period of the control signal S1, thereby acting as an integrator.
In the latter half period of T, the control signal S2 turns on the eighth fet M8 to cause the second constant current source ID to discharge the accumulated charge of the first capacitor C1. During the turn-off period of the control signal S1 and the control signal S2, the first capacitor C1 holds the charge stored in the previous stage, and the second capacitor C2 is continuously discharged through the load resistor. Since the second capacitance C2 is larger than the first capacitance C1, it is ensured that the frequency of the measurement voltage VO has a constant value.
Since the charging and discharging of the first capacitor C1 and the second capacitor C2 are the same in the steady state, the ripple voltage of the measurement voltage VO is proportional to the on-resistance R1, the second capacitor C2 and the resistance RO of the tenth fet M10. Optimizing the on-resistance R1, the second capacitor C2 and the resistance RO of the tenth fet M10 can obtain the ripple voltage of the voltage converter at a lower frequency.
In addition, the ninth fet M9 and the tenth fet M10 are designed to have a lower parasitic capacitance and a higher on-resistance R1, and the on-resistance R1 of the ninth fet M9 and the tenth fet M10 is minimized during charge injection and clock feed-through.
Since the mismatch between the first constant current source IC and the second constant current source ID may cause a deviation of the measurement voltage VO, the first constant current source IC and the second constant current source ID need to be designed with a current mirror circuit to achieve a matching effect.
In addition, the sensing capacitor CS and the delayed feedback output voltage V0 can be approximated by a linear equation, in which the slope of the output voltage V0 with respect to the change of the sensing capacitor CS is proportional to the charging current IC, the intercept of the output voltage V0 depends on the control voltage VC, and the control voltage VC is also a control parameter for calibrating the nominal frequency F0 and the intercept of the output voltage V0 of the voltage-controlled oscillation unit 10. Similarly, the slope of the output voltage V0 resulting from the change in the sense inductor LS is also controlled by the charging current IC, and the control voltage VC adjusts the range of the nominal frequency F0 of the intercept point of the output voltage V0.
In summary, in the present embodiment, frequency modulation is performed before the sensor signal is contaminated by the flicker noise, and the frequency shift caused by the capacitance or inductance change of the sensor is detected through the reactance of the oscillator, so that the advantages of lower phase noise, flicker noise and white noise are achieved at higher frequency, and better noise performance is obtained in output. The logic control circuit adopted by the scheme simulates pulse width modulation configuration and realizes low power consumption, realizes demodulation of frequency modulation signals under the condition that another voltage-controlled oscillator is not needed, and is suitable for corresponding different sensors by controlling the sensitivity, the working range and the nominal point of measurement of the charging current IC and the external voltage VC.
With reference to fig. 1 to 7, the voltage-controlled oscillation unit 10 of the present invention is configured to convert a capacitance variation value Δ C or an inductance variation value Δ L into a frequency variation value Δ F, the waveform conversion unit 20 is configured to convert the frequency variation value Δ F into a time period variation value Δ T, and the frequency-voltage conversion unit 30 is configured to output a measurement voltage VO in a time period variation manner according to the time period variation value Δ T.
The above disclosure is only for the purpose of illustrating the preferred embodiments of the present invention and is not to be construed as limiting the scope of the present invention, therefore, the present invention is not limited by the appended claims.
Claims (10)
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