CN111541387B - CRM boost converter based on variable inductance frequency optimization control - Google Patents

CRM boost converter based on variable inductance frequency optimization control Download PDF

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CN111541387B
CN111541387B CN202010368489.3A CN202010368489A CN111541387B CN 111541387 B CN111541387 B CN 111541387B CN 202010368489 A CN202010368489 A CN 202010368489A CN 111541387 B CN111541387 B CN 111541387B
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resistor
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control
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CN111541387A (en
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刘劲滔
姚凯
刘乐
王泽松
李家镇
高阳
杨坚
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Nanjing University of Science and Technology
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/14—Arrangements for reducing ripples from dc input or output
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4225—Arrangements for improving power factor of AC input using a non-isolated boost converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Abstract

The invention discloses a variable inductance based CRM boost converter with frequency optimization control, which comprises a main power circuit, a CRM control and drive circuit, an output voltage feedback circuit, a multiplier, a rectified input voltage divider circuit and a variable inductance control circuit, wherein the output end of the rectified input voltage divider circuit is connected with the input end of the multiplier, the output end of the output voltage feedback circuit is connected with the other input end of the multiplier, and the output end of the multiplier is connected with one input end of the CRM control and drive circuit; the variable inductance control circuit varies the inductance value of the variable inductance in the main power circuit by applying different bias currents. The invention realizes the reduction of the change range of the switching frequency of the CRM boost PFC converter, improves the efficiency of the converter and can realize the unit power factor.

Description

CRM boost converter based on variable inductance frequency optimization control
Technical Field
The invention relates to an alternating current-direct current converter technology of an electric energy conversion device, in particular to a CRM boost converter based on variable inductance frequency optimization control.
Background
A Power Factor Correction (PFC) converter can reduce input current harmonics and improve an input Power Factor, and has been widely used. The PFC converter is divided into an active mode and a passive mode, and compared with the passive mode, the active mode has the advantages of high input power factor, small size, low cost and the like.
The active PFC converter may employ various circuit topology and control methods, wherein the Boost PFC converter is one of the commonly used PFC converters. The CRM Boost PFC converter is generally applied to medium and small power occasions, and has the advantages that a switching tube is switched on at zero current, a booster diode does not have reverse recovery, PF is high and the like, but the switching frequency of the CRM Boost PFC converter changes along with the change of input voltage and load, and the design of an inductor and an EMI filter is complex.
Aiming at the defect that the variation range of the Switching Frequency of the CRM Boost PFC Converter is large, the YaoKa proposes that the variation range of the Switching Frequency in a wide voltage range is greatly reduced by injecting a proper amount of harmonic waves into input current in a Critical reduction Mode PFC Converter With Fixed Switching Frequency Control document. The harmonic waves of the input current of the method all meet the IEC 61000-3-2Class D standard, and the ratio of the maximum value and the minimum value of the switching frequency is reduced to about 2 times from about 15 times. However, the research method has a complex control circuit, and meanwhile, when the input voltage is higher, the power factor is reduced more obviously, and the harmonic pollution to the power grid is greatly increased.
Disclosure of Invention
The invention aims to provide a CRM boost converter based on variable inductance frequency optimization control, which solves the problem of large switching frequency variation range of a CRM boost PFC converter under the traditional control, improves the efficiency of the converter and can ensure the unit power factor.
The technical solution for realizing the purpose of the invention is as follows: a CRM boost converter based on variable inductance frequency optimization control comprises a main power circuit and a control circuit;
the main power circuit comprises an input voltage source vinEMI filter, diode rectifying circuit RB and variable inductor LbVIAnd a switch tube QbDiode DbFilter capacitor C and load RLd(ii) a Said input voltage source vinThe output cathode of the diode rectifying circuit RB is a reference potential zero point, and the output anode of the diode rectifying circuit RB is connected with the variable inductor LbVIIs connected to one end of a variable inductor LbVIThe other end is respectively connected with a switch tube QbAnd diode DbAnode of (2), diode DbRespectively with one end of the filter capacitor C and the load RLdIs connected with the other end of the filter capacitor C and the load RLdThe other ends of the two ends of the three-phase current transformer are connected with a reference potential zero point and a load RLdThe voltage at both ends is output voltage Vo(ii) a The boost inductor in the main power circuit is a variable inductor LbVIThe switch frequency can be ensured to be always more than 30kHz by applying a fixed bias current to adjust the inductance value to be 0.767mH within the range of 90VAC-110.3VAC of the effective value of the input voltage, applying a fixed bias current to adjust the inductance value to be 1.0304mH within the range of 110.3VAC-249VAC, and applying a fixed bias current to adjust the inductance value to be 0.645mH within the range of 249VAC-264 VAC. The new control not only keeps the advantage that the unit power factor can be realized by the traditional constant on-time control, but also reduces the variation range of the switching frequency by improving the critical inductance value and improves the overall performance of the converter.
The control circuit comprises a CRM control and drive circuit, an output voltage feedback circuit, a rectified input voltage divider circuit, a multiplier and a variable inductance control circuit; the output end of the CRM control and drive circuit and the switching tube QbA gate connection of (a); the input end of the output voltage feedback circuit is connected with the output voltage V of the main power circuitoThe output end of the positive pole of the voltage divider is connected with one input end of the multiplier; input end of rectified input voltage divider circuit and input voltage sampling point VgNamely, the output anode of the diode rectifying circuit RB is connected, and the output end of the diode rectifying circuit RB is connected with the other input end of the multiplier; the output end of the multiplier is connected with one input end of the CRM control and drive circuit; input end of variable inductance control circuit and rectified input voltage sampling point VgI.e. the output anode of the diode rectification circuit RB is connected, the output terminal is connected to the variable inductor LbVIThe above.
Further, the CRM control and drive circuit comprises an inductor LzA sixth resistor RzA seventh resistor RtAn eighth resistor RdZero-crossing detection, RS trigger, drive and first operational amplifier A1；
The inductance LzOne end of the first resistor is connected with a reference point potential zero point, and the other end of the first resistor is connected with a sixth resistor RzOf one terminal of (1), wherein the inductance LzOne end of the reference potential zero point is connected with the variable inductor L in the main power circuitbVIOne end connected with the output anode of the diode rectifying circuit RB is a homonymous end; a sixth resistor RzThe other end of the zero-cross detection circuit is connected with the input end of the zero-cross detection circuit, and the output end of the zero-cross detection circuit is connected with the S end of the RS trigger; the output end of the multiplier is connected with a first operational amplifier A in the CRM control and drive circuit1The non-inverting input terminal of (1); a seventh resistor RtOne end of the switch tube is connected with a reference potential zero point, and the other end of the switch tube is connected with a switch tube QbSource and first operational amplifier a1The first operational amplifier A1The output end of the resistor is connected with the R end of the RS trigger, and the Q end of the RS trigger is driven by an eighth resistor RdAfter being connected in series, the switch tube Q is connectedbA gate electrode of (2).
Further, the output voltage feedback circuit comprises a second operational amplifier A2A third resistor R3A fourth resistor R4A fifth resistor R5And a capacitor C1；
The third resistor R3And an output voltage V of the main power circuitoIs connected to the positive pole of the third resistor R3And the other end of the first resistor and a fourth resistor R4And a second operational amplifier A2Is connected to the reverse input terminal of the fourth resistor R4Is connected to a reference potential zero point, a second operational amplifier A2The positive input terminal of the multiplier is connected with a reference voltage, and the output terminal of the multiplier is connected with one input terminal of the multiplier.
Furthermore, the rectified input voltage divider circuit comprises a first resistor R1And a second resistor R2；
The first resistor R1And a rectified input voltage sampling point VgThat is, the output anode of the diode rectification circuit RB is connected, and the other end of the diode rectification circuit RB is connected with the second resistor R2Is connected to a second resistor R2The other end of the reference point is connected with a reference point zero point.
Further, the multiplier comprises a multiplier;
one input end of the multiplier is connected with the output end of the output voltage feedback circuit, and the other input end of the multiplier is connected with the output end of the input voltage divider circuit.
Further, the variable inductance control circuit comprises a peak sampling chip and a TMS320F28377D chip;
the input end of peak value sampling and the sampling point V of rectified input voltagegNamely, the output anode of the diode rectifying circuit RB is connected, the output end of the diode rectifying circuit RB is connected with the ADC input end of the TMS320F28377D chip, and the DAC1 output port of the TMS320F28377D chip is connected with the variable inductor LbVIAre connected.
Compared with the prior art, the invention has the remarkable advantages that: (1) under control, the power factor of the converter is still 1, and the control circuit is simple; (2) except that the critical inductance value is the same as that of the traditional control when the voltage is 264VAC, the change range of the switching frequency can be greatly reduced under other input voltages through the variable inductance technology; (3) the lowest frequency in the wide voltage range is constant at 30 kHz; (4) the design of the EMI filter and the inductor is simplified, the input filtering effect is improved, the switching loss and the magnetic core loss are reduced, and the efficiency of the converter is improved.
Drawings
Fig. 1 is a schematic circuit diagram of a CRM boost converter based on frequency optimization control of variable inductance in an embodiment of the present invention.
Fig. 2 is a schematic diagram of a main circuit of a Boost PFC converter in an embodiment of the present invention.
Fig. 3 is a graph of the inductor current waveform of the CRM Boost PFC converter in an embodiment of the present invention.
Fig. 4 is a waveform diagram of the inductor current of the CRM Boost PFC converter within half the power frequency cycle in an embodiment of the present invention.
Fig. 5 is a graph showing the inductance value variation over a wide input voltage range according to the embodiment of the present invention.
Fig. 6 is a graph of minimum switching frequency variation over a wide input voltage range in accordance with an embodiment of the present invention.
Fig. 7 is a graph of minimum switching frequency variation under conventional control combined with variable inductance control in an embodiment of the present invention.
Fig. 8 is a graph showing a variation of a minimum switching frequency under frequency optimization control based on a variable inductance according to an embodiment of the present invention.
FIG. 9 shows a variable inductor L in the main power circuit according to an embodiment of the present inventionbVIA basic model diagram of (2).
Fig. 10 is a graph of the variation of the critical inductance at different input voltages according to an embodiment of the present invention.
FIG. 11 is a graph showing the minimum value of the switching frequency in a wide input voltage range under two kinds of control in the embodiment of the present invention.
Fig. 12 is a graph of the variation of the switching frequency in the second half of the power frequency cycle under the two controls in the embodiment of the present invention.
Detailed Description
The invention is described in further detail below with reference to the figures and the embodiments.
Working principle of 1 CRM Boost PFC converter
Fig. 2 is a Boost PFC converter main circuit.
Setting: 1. all devices are ideal elements; 2. the output voltage ripple is very small compared to its dc amount;
3. the switching frequency is much higher than the input voltage frequency.
Fig. 3 is a waveform of the inductor current in one switching cycle at CRM. When Q isbWhen conducting, DbCut-off and boost inductor LbVoltage across vgInductor current iLbStarting from zero with vg/LbThe slope of (a) rises linearly; when Q isbWhen turned off, iLbBy DbFollow current, at this time LbVoltage across vg－Vo，iLbWith (V)o－vg)/LbSince the Boost converter operates in CRM mode, at iLbWhen the voltage drops to zero, the switch tube QbOn, a new switching cycle is started.
Without loss of generality, define the input AC voltage vinThe expression of (a) is:
vin＝Vm sin ωt (1)
wherein VmAnd ω is the amplitude and angular frequency of the input ac voltage, respectively;
then the rectified voltage v of the input voltagegComprises the following steps:
vg＝Vm·|sin ωt| (2)
peak value i of inductor current in one switching periodLb_pkComprises the following steps:
Figure BDA0002477311280000041
wherein t isonIs QbOn-time of (d);
in each switching cycle, the boost inductor LbVolt-second area balance at both ends, so QbOff time t ofoffComprises the following steps:
Figure BDA0002477311280000051
as can be seen from fig. 3, the average value i of the inductor current in each switching cycleLb_avHalf of its peak value, which can be obtained from equation (3):
Figure BDA0002477311280000052
as can be seen from equation (5), if within one power frequency period, the on-time tonIs fixed, the average value of the inductor current is sinusoidal, i.e. the input power factor is 1. As can be seen from equation (4), the off time toffThe switching frequency is changed along with the change of the input voltage instantaneous value, namely the switching frequency is changed continuously in a power frequency period.
Fig. 4 is a graph of the waveforms of the inductor current, the peak envelope and the mean value over half the power frequency period.
As can be seen from equation (5) and FIG. 2, the input current iinComprises the following steps:
Figure BDA0002477311280000053
setting the output power of the converter to PoThe efficiency is 1, and the input and output power balance can be obtained:
Figure BDA0002477311280000054
the on-time t can be obtained from the equation (7)onComprises the following steps:
Figure BDA0002477311280000055
by substituting the formula (8) for the formulae (5) and (6), the average value i of the inductor current can be obtainedLb_avAnd an input current iinComprises the following steps:
Figure BDA0002477311280000056
Figure BDA0002477311280000057
wherein 2Po/VmIs the fundamental current amplitude;
the turn-off time t is obtained from the equations (4) and (8)offComprises the following steps:
Figure BDA0002477311280000061
the switching frequency f can be obtained by combining the formula (8) and the formula (11)sComprises the following steps:
Figure BDA0002477311280000062
the above formula can be:
Figure BDA0002477311280000063
by observing the formula (13), it can be found that the switching frequency f in the power frequency period is changed along with the constant change of ω t when the converter parameter is determinedsAnd accordingly, is constantly changing. The switching frequency is [0, π/2]Within the interval, the value decreases monotonically with ω t and is [ pi/2, pi ]]Within the interval, the interval monotonically increases with ω t, so that the minimum value f within the power frequency period iss_minAnd maximum value fs_maxOccurring at the peak time and zero-crossing time of the input voltage, respectively, i.e. at a minimum value when ω t is pi/2 and at a maximum value when ω t is 0 or pi, i.e. at
Figure BDA0002477311280000064
Figure BDA0002477311280000065
According to formulae (14) and (15), with fs_maxRatio fs_minCan obtain the product
Figure BDA0002477311280000066
If the lowest switching frequency is limited to 30kHz, the maximum inductance value L can be obtained from the equation (12)b_maxIs expressed as
Figure BDA0002477311280000067
According to equation (18), a variation curve of the critical inductance value in a wide input voltage range can be obtained by combining specific parameters of the converter, as shown in fig. 5 below.
2 realizing frequency optimization control strategy based on variable inductance
The frequency optimization control provided by the invention utilizes a variable inductance technology to solve the problem of large switching frequency variation range under the control of the traditional fixed conduction time. Under the frequency optimization control based on the variable inductor, when a critical inductance value is designed, the critical inductance value is not limited by the minimum switching frequency any more, a larger inductance value can be selected, and when the switching frequency is lower than the audible frequency of 30kHz in a wide input voltage range, the value of the variable inductor is adjusted by applying bias current to the variable inductor control winding, so that the switching frequency is increased, and the requirement of the minimum switching frequency is met. Therefore, the critical inductance value is improved, the switching frequency variation range is reduced, the switching frequency in a wide input voltage range can be kept to be larger than 30kHz, and the design requirement of the converter is met.
According to the idea of frequency optimization control and analysis in conjunction with fig. 6, when the selected inductance value is greater than the minimum critical inductance value, the minimum value of the switching frequency is first lower than 30kHz at both sides of the low voltage and the high voltage, at this time, two voltage nodes whose inductance values need to be adjusted exist, which are called as adjustment points of the variable inductor, and the switching frequency can be increased to more than 30kHz only by adjusting the value of the inductor at the adjustment points of the variable inductor, as shown in fig. 7, the minimum switching frequency variation curve under the control of the conventional control in conjunction with the variable inductor is shown.
In combination with the above analysis, the frequency optimization control performs inductance value conversion only at two variable inductance adjustment points in order to simplify the control circuit as much as possible. In a certain range of low-voltage side, a fixed bias current i is applied to the variable inductance control windingbias1Reducing the inductance value to Vm_rmsThe corresponding inductance critical value at 90V is 0.767 mH; within a certain range of the high-voltage side, a fixed bias current i is applied to the variable inductance control windingbias2Reducing the inductance value to Vm_rmsThe corresponding inductance threshold at 264V is 0.645 mH. As can be seen from fig. 7, the new control method selects a larger inductance value, and the portions of the two sides with the switching frequency lower than 30kHz will be influenced by the inductance value adjustment and move upward as a whole, so that there must exist an optimal inductance value and two optimal adjustment points of the variable inductor, so that the variation range of the switching frequency within the wide input voltage range is minimized. The following solution of the optimal inductance value and two optimal adjustment points of the variable inductance is performed：
Writing equation (12) as relating to inductance LbAnd an input voltage VmIs shown in formula (19)
Figure BDA0002477311280000071
The observation (19) is a function of the input voltage VmThe cubic function can be calculated to obtain the maximum value point of the minimum switching frequency curve in V within a wide input voltage rangemTaken at 267 VAC. Meanwhile, the change rate of the analysis formula (19) on the two sides of the maximum point can be obtained, and the change rate of the analysis formula (19) on the high-voltage side is higher, so that the change range of the switching frequency on the high-voltage side becomes a key factor for limiting the change range of the minimum value of the switching frequency.
Assume an optimal inductance value of LoptimalThe optimal adjustment point of the variable inductance on the high-voltage side is Vh_optimalThe optimal adjustment point of the variable inductance on the low-voltage side is Vl_optimalEquation of availability
Figure BDA0002477311280000081
In combination with the specific parameters of the converter, L can be solvedoptimal＝1030.4uH,Vh_optimal＝352.14VAC,Vl_optimal155.95VAC, and the minimum switching frequency variation range is 30 kHz-47.9 kHz.
According to the calculated optimal inductance value LoptimalOptimum adjustment point V of variable inductance on high-voltage sideh_optimal_rms249V, low-voltage side variable inductance optimal regulation point Vl_optimal_rms110.3V, in combination with equation (19), a variation curve of the minimum switching frequency under the frequency optimization control based on the variable inductance can be drawn, as shown in fig. 8. As can be seen from FIG. 8, the inductance value of the variable inductor can be adjusted twice within a wide input voltage range, i.e., the inductance value is adjusted to 0.767mH by passing a fixed bias current within the effective value range of the input voltage of 90VAC-110.3VAC, 1.0304mH is not passed within the range of 110.3VAC-249VAC, and 249VAC-26The inductance value is adjusted to 0.645mH by passing a fixed bias current in the range of 4VAC, and the switching frequency is always more than 30 kHz. The new control not only keeps the advantage that the unit power factor can be realized by the traditional constant on-time control, but also reduces the variation range of the switching frequency by improving the critical inductance value and improves the overall performance of the converter.
The basic model of variable inductance is shown in fig. 9, and is composed of two side auxiliary windings and a middle main winding, and the auxiliary windings N are controlled to flow throughCBias current of (I)biasCan change the inductance L of the main magnetic corebVIIn the present invention, a double E-type core is used, as shown in fig. 9. Main induction winding NLWound on a central core with an air gap, auxiliary winding NCWound on two side cores and two auxiliary windings connected in series to eliminate the current I from the main inductorLbVIInduced voltage due to ripple. When no bias current exists, the main winding maintains the initial inductance value which is the same as the normal inductance; when there is a bias current IbiasFlows through NCThen, a bias flux phi is generated along the external path of the double E-shaped magnetic corebiasWith phibiasIncrease of phibiasThe working point of the magnetic core on the B-H curve is pushed from the linear region to the nonlinear saturation region, the magnetic permeability mu of the magnetic core along the path is reduced, and when the main winding is electrified, main magnetic flux phi can be generatedmainDue to main magnetic flux phimainThe main magnetic core is also affected by the bias current and the magnetic permeability is reduced by flowing through the middle magnetic core and the external path. To sum up, IbiasThe effective permeability on the intermediate core is reduced, resulting in a main inductance LbVIAnd decreases.
From the basic model of variable inductance of fig. 9, the calculation formula of variable inductance can be derived as:
Figure BDA0002477311280000082
in the formula I1，l3，lgThe lengths of the auxiliary winding, the main winding and the air gap effective magnetic circuit are respectively; a. the1、A3Is assisted byThe effective cross-sectional areas of the magnetic core and the main magnetic core; n is3Is the number of turns of the main winding; mu.s0Is the air permeability; mu.s3And muvarThe effective permeability of the main and auxiliary windings respectively.
As can be seen from equation (21), the variable inductance is substantially a change in μ by the bias current3And muvarI.e. the effective permeability of the main and auxiliary windings.
A variable inductance model is built in simulation software LTSPICE, and variable inductance L is drawnbVIWith the inductance value of the bias current IbiasThe variation is shown in fig. 9. The variable inductance parameter designed by the invention is combined, namely the inductance value is adjusted to 0.767mH by applying the fixed bias current within the range of 90VAC-110.3VAC of the effective value of the input voltage, the inductance value is not adjusted to 1.0304mH by applying the bias current within the range of 110.3VAC-249VAC, and the inductance value is adjusted to 0.645mH by applying the fixed bias current within the range of 249VAC-264 VAC.
3 comparison of Performance
3.1 critical inductance value
As can be seen from the analysis of fig. 5 and 10, under the conventional constant on-time control, in order to ensure that the switching frequencies are all greater than the lowest frequency of 30kHz within the wide input voltage range, the critical inductance value is limited by the minimum value of 264VAC of the input voltage, which is only designed to be 0.645 mH. However, under the frequency-optimized control based on the variable inductor, the value of the inductance is decreased when the bias current is increased according to the characteristics of the variable inductor, and the threshold inductance is selected without being limited by the minimum value of the inductor in a wide input voltage range. Therefore, the optimal inductance L calculated in the previous step can be selected when designing the critical inductanceoptimal1030.4 uH. Compared with the traditional constant on-time control, the critical inductance value under the frequency optimization control based on the variable inductor is obviously improved.
3.2 variation of switching frequency
According to equation (19) and in combination with the foregoing analysis, the minimum variation curve of the switching frequency in a wide input voltage range under two controls can be plotted, as shown in fig. 11. It can be seen from fig. 11 that, after the new frequency optimization control based on variable inductance is adopted, the change range of the minimum value of the switching frequency is reduced from 30kHz to 78kHz under the traditional control to 30kHz to 47.9kHz, and the change range of the switching frequency is greatly reduced.
Further analyzing equation (12), and bringing the critical inductance values 0.645mH and 1.0304mH under the conventional constant on-time control and variable-inductor-based frequency optimization control into equation (12), respectively, a variation curve of the switching frequency in the half power frequency period under the two controls can be made, as shown in FIG. 12, where fs1(Vm) The corresponding solid line is the frequency curve under conventional control, fs2(Vm) The corresponding dotted line is the frequency curve under variable inductance frequency optimization control. As can be seen from the observation of FIG. 12, the variation range of the switching frequency is reduced in different degrees under different input voltages, and under the most common standard input voltages of 110VAC and 220VAC, the variation range of the switching frequency is respectively reduced from 47 kHz-79 kHz and 70 kHz-315 kHz under the traditional control to 30 kHz-49 kHz and 43 kHz-194 kHz under the variable inductance frequency optimization control, and the switching frequency is greatly reduced as a whole. The change range of the traditional control switch frequency at the low-voltage 90VAC is smaller, but the new control can further narrow the change range to 30 kHz-44 kHz; at the high voltage 264VAC, the switching frequency variation curves are completely overlapped in FIG. 12 because the new control method maintains the same critical inductance value as the conventional control. In summary, the new control can effectively reduce the maximum switching frequency f simultaneouslys_maxAnd minimum switching frequency fs_minIs such that the absolute value | f of the switching frequency variation ranges_max-fs_minThe | is reduced, but the ratio of the two is always kept constant.
4 CRM boost converter based on variable inductance frequency optimization control
With reference to fig. 1, the rectified input voltage vgThrough a first resistor R1And a second resistor R2Partial pressure to obtain vA＝kvgVmL sin ω t l, where kvgIs the coefficient of partial pressure, kvg＝R2/(R1+R2) (ii) a The output voltage passes through a third resistor R3And a fourth resistor R4To obtain a divided voltage vB＝kvgVoWherein R is3/R4＝R1/R2。
Voltage division v in voltage ring control circuitBAnd reference voltage V of error amplifierrefIn comparison, where Vref2.5V via a fifth resistor R5And a capacitor C1The constituent regulator deriving an error signal vEA，vEAAnd vAThe point voltage v is obtained after being connected into a multiplierEComprises the following steps:
vE＝vEAkvgVm|sinωt| (22)
voltage v of equation (22)EAnd a seventh resistor RtControl switch tube Q after voltage comparisonbTurn off, sixth resistor RzThe voltage on the switch tube Q is controlled after zero detectionbThe on-time of the change rule shown in the formula (8) is obtained.
With reference to fig. 1, the CRM boost converter for frequency optimization control based on variable inductance of the present invention includes a main power circuit and a control circuit;
the main power circuit comprises an input voltage source vinEMI filter, diode rectifying circuit RB and variable inductor LbVIAnd a switching tube QbDiode DbFilter capacitor C and load RLd(ii) a Said input voltage source vinThe output cathode of the diode rectifying circuit RB is a reference potential zero point, and the output anode of the diode rectifying circuit RB is connected with the variable inductor LbVIIs connected to one end of a variable inductor LbVIThe other end is respectively connected with a switch tube QbAnd diode DbAnode of (2), diode DbRespectively connected with one end of the filter capacitor C and the load RLdIs connected to the other end of the filter capacitor C and the load RLdThe other ends of the two ends of the three-phase current transformer are connected with a reference potential zero point and a load RLdThe voltage at both ends is output voltage Vo(ii) a The boost inductor in the main power circuit is a variable inductor LbVIAt the input voltage effective value of 90VAC-110.The switch frequency can be ensured to be always more than 30kHz by switching on the fixed bias current within the range of 3VAC to adjust the inductance value to be 0.767mH, switching off the bias current within the range of 110.3VAC-249VAC to adjust the inductance value to be 1.0304mH, and switching on the fixed bias current within the range of 249VAC-264VAC to adjust the inductance value to be 0.645 mH. The new control not only keeps the advantage that the unit power factor can be realized by the traditional constant on-time control, but also reduces the variation range of the switching frequency by improving the critical inductance value and improves the overall performance of the converter.
The control circuit comprises a CRM control and drive circuit, an output voltage feedback circuit, a rectified input voltage divider circuit, a multiplier and a variable inductance control circuit; the output end of the CRM control and drive circuit and the switching tube QbA gate connection of (a); the input end of the output voltage feedback circuit is connected with the output voltage V of the main power circuitoThe output end of the positive pole of the voltage divider is connected with one input end of the multiplier; input end of rectified input voltage divider circuit and input voltage sampling point VgNamely, the output anode of the diode rectifying circuit RB is connected, and the output end of the diode rectifying circuit RB is connected with the other input end of the multiplier; the output end of the multiplier is connected with one input end of the CRM control and drive circuit; input end of variable inductance control circuit and rectified input voltage sampling point VgNamely, the output anode of the diode rectifying circuit RB is connected, and the output terminal is connected to the variable inductor.
Further, the CRM control and drive circuit 2 comprises an inductor LzA sixth resistor RzA seventh resistor RtAn eighth resistor RdZero-crossing detection, RS trigger, drive and first operational amplifier A1；
The inductance LzOne end of the first resistor is connected with a reference point potential zero point, and the other end of the first resistor is connected with a sixth resistor RzOf one terminal of (1), wherein the inductance LzOne end of the reference potential zero point is connected with the variable inductor L in the main power circuitbVIOne end connected with the output anode of the diode rectifying circuit RB is a homonymous end; a sixth resistor RzThe other end of the zero-cross detection circuit is connected with the input end of the zero-cross detection circuit, and the output end of the zero-cross detection circuit is connected with the S end of the RS trigger; the output end of the multiplier is connected with the CRM control and driveA first operational amplifier A in the circuit1The non-inverting input terminal of (1); a seventh resistor RtOne end of the switch tube is connected with a reference potential zero point, and the other end of the switch tube is connected with a switch tube QbSource and first operational amplifier a1The first operational amplifier A1The output end of the resistor is connected with the R end of the RS trigger, and the Q end of the RS trigger is driven by an eighth resistor RdAfter being connected in series, the switch tube Q is connectedbA gate electrode of (a).
Further, the output voltage feedback circuit 3 includes a second operational amplifier A2A third resistor R3A fourth resistor R4A fifth resistor R5And a capacitor C1；
The third resistor R3And the output voltage V of the main power circuitoIs connected to the positive pole of the third resistor R3The other end of (2) and a fourth resistor R4And a second operational amplifier A2Is connected to the reverse input terminal of the fourth resistor R4Is connected to a reference potential zero point, a second operational amplifier A2The positive input terminal of the multiplier is connected with a reference voltage, and the output terminal of the multiplier is connected with one input terminal of the multiplier.
Further, the input voltage divider circuit 4 includes a first resistor R1And a second resistor R2；
The first resistor R1One end of and an input voltage sampling point VgThat is, the output anode of the diode rectification circuit RB is connected, and the other end of the diode rectification circuit RB is connected with the second resistor R2Is connected to a second resistor R2The other end of the reference point is connected with a reference point zero point.
Further, the multiplier 5 includes a multiplier;
one input end of the multiplier is connected with the output end of the output voltage feedback circuit, and the other input end of the multiplier is connected with the output end of the rectified input voltage divider circuit.
Further, the variable inductance control circuit comprises a peak sampling chip and a TMS320F28377D chip;
the input end of the peak value sampling and the rectified input voltage samplingSampling point VgNamely, the output anode of the diode rectifying circuit RB is connected, the output end of the diode rectifying circuit RB is connected with the ADC input end of the TMS320F28377D chip, and the DAC1 output port of the TMS320F28377D chip is connected with the variable inductor LbVIAre connected.

Claims (6)

1. A CRM boost converter based on variable inductance frequency optimization control is characterized by comprising a main power circuit (1) and a control circuit;
the main power circuit (1) comprises an input voltage sourcev in EMI filter, diode rectifier circuitRB、Variable inductanceL bVI Switch tubeQ b Diode, and method for manufacturing the sameD b Filter capacitorCAnd a loadR Ld (ii) a The input voltage sourcev in Connected with the input port of the EMI filter, the output port of the EMI filter is connected with the diode rectifying circuitRBIs connected with the input port of the diode rectifying circuitRBThe output cathode of the rectifier is a reference potential zero point, and the diode rectifier circuitRBOutput positive electrode and variable inductorL bVI Is connected to one end of a variable inductorL bVI The other end is respectively connected with a switch tubeQ b Drain electrode and diodeD b Anode of (2), diodeD b Respectively with a filter capacitorCAnd a loadR Ld Is connected to a filter capacitorCAnother end and a loadR Ld The other ends of the two ends of the three-phase current transformer are connected with a reference potential zero point and a loadR Ld The voltage at both ends is the output voltageV o Output voltageV o The anode of the voltage feedback circuit is connected with the input end of the output voltage feedback circuit (3); the boost inductor in the main power circuit (1) is a variable inductorL bVI The inductance value is adjusted to an optimal value by applying bias current under different input voltages, the fixed bias current is introduced within the range of 90VAC-110.3VAC of the effective value of the input voltage to adjust the inductance value to 0.767mH, the bias current is not introduced within the range of 110.3VAC-249VAC, the inductance value is 1.0304mH, and the bias current is not introduced within the range of 249VACThe inductance value is adjusted to 0.645mH by passing a fixed bias current in the VAC-264VAC range, so that the switching frequency is always higher than 30 kHz;
the control circuit comprises a CRM control and drive circuit (2), an output voltage feedback circuit (3), a rectified input voltage divider circuit (4), a multiplier (5) and a variable inductance control circuit (6); the output end of the CRM control and drive circuit (2) and a switch tubeQ b A gate connection of (a); the input end of the output voltage feedback circuit (3) is connected with the output voltage of the main power circuit (1)V o The output end of the positive pole of the voltage regulator is connected with one input end of the multiplier (5); the input end of the rectified input voltage divider circuit (4) and the sampling point of the rectified input voltageV g I.e. diode rectifier circuitRBThe output positive pole of the multiplier (5) is connected, and the output end of the multiplier is connected with the other input end of the multiplier; the output end of the multiplier (5) is connected with one input end of the CRM control and drive circuit (2) and a switching tubeQ b The source of the CRM control and drive circuit (2) is connected with the other input end of the CRM control and drive circuit; input end of variable inductance control circuit (6) and rectified input voltage sampling pointV g I.e. diode rectifier circuitRBIs connected to the output positive pole, the output terminal is connected to the variable inductance of the main power circuit (1)L bVI On the control terminal of (c).
2. A CRM boost converter based on variable inductance frequency optimization control according to claim 1, characterized in that, the CRM control and drive circuit (2) comprises an inductanceL z And a sixth resistorR z A seventh resistorR t An eighth resistorR d Zero-crossing detection, RS trigger, drive and first operational amplifierA 1；
The inductorL z One end of the first resistor is connected with a reference potential zero point, and the other end of the first resistor is connected with a sixth resistorR z One terminal of (1), wherein the inductorL z One end of the reference potential zero point is connected with the variable inductor in the main power circuit (1)L bVI Connecting diodeRectifying circuitRBOne end of the output anode is a homonymous end; sixth resistorR z The other end of the zero-cross detection circuit is connected with the input end of the zero-cross detection circuit, and the output end of the zero-cross detection circuit is connected with the S end of the RS trigger; the output end of the multiplier (5) is connected with a first operational amplifier in the CRM control and drive circuit (2)A 1The non-inverting input terminal of (1); seventh resistorR t One end of the switch tube is connected with a reference potential zero point, and the other end of the switch tube is connected with the switch tubeQ b Source and first operational amplifierA 1The inverting input terminal of, the first operational amplifierA 1The output end of the resistor is connected with the R end of the RS trigger, and the Q end of the RS trigger is driven by the eighth resistorR d After being connected in series, the switch tube is connectedQ b A gate electrode of (2).
3. The CRM boost converter based on variable inductance frequency optimization control of claim 1, wherein the output voltage feedback circuit (3) comprises a second operational amplifierA 2A third resistorR 3A fourth resistorR 4A fifth resistorR 5And a capacitorC 1；
The third resistorR 3And the output voltage of the main power circuit (1)V o Is connected to the positive pole of the third resistorR 3The other end of the resistor and a fourth resistorR 4And a second operational amplifierA 2Is connected to the reverse input terminal of the fourth resistorR 4Is connected to a reference potential zero point, a second operational amplifierA 2The positive input terminal of the multiplier (5) is connected with a reference voltage, and the output terminal of the multiplier is connected with one input terminal of the multiplier (5).
4. The CRM boost converter based on variable inductance frequency optimization control of claim 1, wherein the input voltage divider circuit (4) comprises a first resistorR 1And a second resistorR 2；
The first resistorR 1One terminal of and an input voltage sampling pointV g I.e. diode rectifier circuitRBIs connected with the output anode, and the other end of the output anode is connected with a second resistorR 2Is connected to a second resistorR 2The other end of the reference point is connected with a reference point zero point.
5. A CRM boost converter for variable inductance based frequency optimized control according to claim 1, characterized in that said multiplier (5) comprises a multiplier;
one input end of the multiplier is connected with the output end of the output voltage feedback circuit (3), and the other input end of the multiplier is connected with the output end of the input voltage divider circuit (4).
6. The CRM boost converter based on variable inductance frequency optimization control of claim 1, characterized in that, the variable inductance control circuit (6) comprises a peak sampling, TMS320F28377D chip;
the input end of peak value sampling and the sampling point of rectified input voltageV g I.e. diode rectifier circuitRBThe output anode of the TMS320F28377D chip is connected, the output end of the TMS320F28377D chip is connected with the ADC input end of the chip, and the DAC1 output port of the TMS320F28377D chip is connected with the variable inductorL bVI And (4) connecting.
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Publication number Priority date Publication date Assignee Title
EP2086095A1 (en) * 2008-10-15 2009-08-05 China Electronics Weihua Co., Ltd. Phase shift and synchronization control circuit for interleaved PFC controller
CN108696158A (en) * 2018-06-22 2018-10-23 南京航空航天大学 The compensating control method of CRM Boost pfc converter input filter capacitor electric currents
CN110829822A (en) * 2018-08-07 2020-02-21 南京理工大学 CRM Boost PFC converter for optimizing frequency variation range
CN110829827A (en) * 2018-08-07 2020-02-21 南京理工大学 CRM boost-buck PFC converter with constant switching frequency

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP2086095A1 (en) * 2008-10-15 2009-08-05 China Electronics Weihua Co., Ltd. Phase shift and synchronization control circuit for interleaved PFC controller
CN108696158A (en) * 2018-06-22 2018-10-23 南京航空航天大学 The compensating control method of CRM Boost pfc converter input filter capacitor electric currents
CN110829822A (en) * 2018-08-07 2020-02-21 南京理工大学 CRM Boost PFC converter for optimizing frequency variation range
CN110829827A (en) * 2018-08-07 2020-02-21 南京理工大学 CRM boost-buck PFC converter with constant switching frequency

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