CN110542887A - linear frequency modulation signal carrier suppression method for radar system - Google Patents

linear frequency modulation signal carrier suppression method for radar system Download PDF

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Publication number
CN110542887A
CN110542887A CN201910784554.8A CN201910784554A CN110542887A CN 110542887 A CN110542887 A CN 110542887A CN 201910784554 A CN201910784554 A CN 201910784554A CN 110542887 A CN110542887 A CN 110542887A
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voltage
output
mixer
direct current
bias voltage
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CN110542887B (en
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岳义崴
沙瑜
王宇
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Institute of Electronics of CAS
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Institute of Electronics of CAS
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    • GPHYSICS
    • G01MEASURING; TESTING
    • G01SRADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88Radar or analogous systems specially adapted for specific applications
    • G01S13/89Radar or analogous systems specially adapted for specific applications for mapping or imaging
    • G01S13/90Radar or analogous systems specially adapted for specific applications for mapping or imaging using synthetic aperture techniques, e.g. synthetic aperture radar [SAR] techniques
    • GPHYSICS
    • G01MEASURING; TESTING
    • G01SRADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/36Means for anti-jamming, e.g. ECCM, i.e. electronic counter-counter measures

Abstract

the embodiment of the application discloses a method for suppressing a linear frequency modulation signal carrier wave of a radar system, which comprises the following steps: applying a first direct current bias voltage to an I-path baseband linear frequency modulation signal sent by an I-path output end of the digital baseband circuit; applying a second direct current bias voltage to a Q-path baseband linear frequency modulation signal sent by a Q-path output end of the digital baseband circuit; and the optimization of the carrier suppression effect is realized by sequentially adjusting the first direct current bias voltage to a first voltage value corresponding to the lowest value of the carrier power output by the signal modulation system and the second direct current bias voltage value corresponding to a second voltage value corresponding to the lowest value of the carrier power output by the signal modulation system.

Description

Linear frequency modulation signal carrier suppression method for radar system
Technical Field
the invention relates to the technical field of synthetic aperture linear frequency modulation signal generation, in particular to a radar system linear frequency modulation signal carrier suppression method.
background
in a high-resolution satellite-borne synthetic aperture radar system, the carrier suppression degree of a chirp signal is an important index, and if the carrier energy of the chirp signal transmitted by the system is too high, the pulse response integral sidelobe ratio generated after the system is subjected to pulse compression is deteriorated, so that the imaging quality is influenced. At present, the main methods for inhibiting the linear frequency modulation signal carrier wave of the satellite-borne synthetic aperture radar are as follows: firstly, when hardware circuits of In-phase (I) path and Quadrature (Q) path linear frequency modulation signals are designed, direct current bias voltages on the two paths of baseband linear frequency modulation signals are kept consistent as much as possible, and when the direct current bias voltages on the two paths of baseband linear frequency modulation signals are consistent, theoretical analysis shows that the Quadrature-modulated signals do not contain carrier components; secondly, in the quadrature modulation circuit, two used mixers or modulators are screened, and devices with small carrier leakage are selected as much as possible, and meanwhile, IF a direct current bias voltage exists at an Intermediate Frequency (IF) port of the mixer, the energy of a carrier (LO) leaked in a Radio Frequency (RF) port is affected.
the above two methods for suppressing the carrier of the chirp signal of the radar system cannot unify the carrier suppression effects of the digital baseband circuit and the analog quadrature modulation circuit, and cannot compensate each other to achieve the minimum carrier energy.
disclosure of Invention
in view of this, the present application provides a method for suppressing a chirp signal carrier of a radar system.
the embodiment of the application provides a method for suppressing a linear frequency modulation signal carrier of a radar system, which is applied to a signal modulation system, wherein the signal modulation system comprises: the system comprises a digital baseband circuit, a first low-pass filter, a first frequency mixer, a second low-pass filter, a second frequency mixer, a local oscillation source, a 90-degree power divider and a 0-degree power combiner; wherein the digital baseband circuit has an in-phase I-path output end and a quadrature Q-path output end, the digital baseband circuit is connected to the input end of the first low-pass filter through the I-path output end, the digital baseband circuit is connected to the input end of the second low-pass filter through the Q-path output end, the output end of the first low-pass filter is connected to the intermediate frequency IF end of the first mixer, the radio frequency RF end of the first mixer is connected to the first input end of the 0 ° power combiner, the output end of the second low-pass filter is connected to the IF end of the second mixer, the RF end of the second mixer is connected to the second input end of the 0 ° power combiner, the local oscillator source is connected to the input end of the 90 ° power divider, and the first output end of the 90 ° power divider is connected to the carrier LO end of the first mixer, a second output end of the 90 ° power divider is connected to an LO end of the second mixer;
the method comprises the following steps:
Applying a first direct current bias voltage to the I-path linear frequency modulation signal sent by the I-path output end;
Applying a second direct current bias voltage to the Q-path baseband linear frequency modulation signal sent by the Q-path output end;
adjusting the voltage value of the first direct current bias voltage in a first preset voltage interval, and detecting a first carrier power change curve of carrier power output by the output end of the 0-degree power combiner, which corresponds to the change of the voltage value of the first direct current bias voltage in the first preset voltage interval;
Taking the voltage value of the first direct current bias voltage corresponding to the lowest value of the carrier power on the first carrier power variation curve as a first voltage value;
adjusting the first DC bias voltage to the first voltage value;
adjusting the voltage value of the second direct current bias voltage within a second preset voltage interval, and detecting a second carrier power change curve of the carrier power output by the output end of the 0-degree power combiner, which corresponds to the change of the voltage value of the second direct current bias voltage within the second preset voltage interval;
Taking the voltage value of the second direct current bias voltage corresponding to the lowest carrier power value on the second carrier power variation curve as a second voltage value;
and adjusting the second direct current bias voltage to the second voltage value.
in the above technical solution, the applying a first dc bias voltage to the I-band chirp signal sent by the I-band output terminal includes:
The positive pole of the first direct current voltage is connected with the I-path output end through a first resistor R1, and the negative pole of the first direct current voltage is connected with the I-path output end through a second resistor R2; the positive electrode potential of the first direct-current voltage is + V1, the negative electrode potential of the first direct-current voltage is-V1, and V1 is greater than 0;
The expression of the first dc bias voltage V α is:
the expression of the I-baseband linear frequency modulation signal fI is fI ═ cos (pi kt2), the expression of the local oscillation frequency f η output by the local oscillation source is f η ═ sin (ω t), wherein k is the frequency modulation slope, t is time, and ω is angular frequency;
the expression of the signal fIRF output by the RF terminal of the first mixer after passing through the first mixer is:
f=[cos(πkt)+V]·sin(ωt);
The LO end of the first mixer leaks a signal to the RF end of the first mixer, wherein the signal is p times of f eta, and p is greater than 0;
Then, the expression of the carrier f σ output by the RF end of the first mixer is:
in the above technical solution, the applying a second dc bias voltage to the Q-band chirp signal sent by the Q-band output terminal includes:
The positive pole of the second direct current voltage is connected with the Q-path output end through a third resistor R3, and the negative pole of the second direct current voltage is connected with the Q-path output end through a fourth resistor R4; the positive electrode potential of the second direct-current voltage is + V2, the negative electrode potential of the second direct-current voltage is-V2, and V2 is greater than 0;
the expression of the second dc bias voltage V β is:
the expression of the Q baseband linear frequency modulation signal fQ is fQ ═ sin (pi kt 2);
the expression of the signal fQRF outputted by the RF end of the second mixer after fQ passes through the second mixer is:
f=[sin(πkt)+V]·sin(ωt);
the LO end of the second mixer leaks a signal of the RF end of the second mixer to be q times of f eta, and q is larger than 0;
then, the expression of the carrier f σ output by the RF end of the second mixer is:
in the above technical solution, the adjusting the voltage value of the first dc bias voltage within a first preset voltage interval includes:
Adjusting V1 and/or R1 and/or R2 so that V α varies within the first preset voltage interval.
in the above technical solution, the adjusting the voltage value of the second dc bias voltage within the second preset voltage interval includes:
adjusting V2 and/or R3 and/or R4 so that V β varies within the second preset voltage interval.
according to the method for suppressing the carrier wave of the linear frequency modulation signal of the radar system, a first direct current bias voltage is applied to the linear frequency modulation signal of the I-path baseband sent by the I-path output end of a digital baseband circuit; applying a second direct current bias voltage to a Q-path baseband linear frequency modulation signal sent by a Q-path output end of the digital baseband circuit; adjusting the first direct current bias voltage to a first voltage value corresponding to the lowest value of the carrier power output by the signal modulation system and adjusting the second direct current bias voltage value to a second voltage value corresponding to the lowest value of the carrier power output by the system; the carrier suppression effect of the digital baseband circuit and the quadrature modulation circuit is combined, and the carrier suppression degree is improved.
Drawings
the drawings illustrate generally, by way of example, but not by way of limitation, various embodiments discussed herein.
Fig. 1 is a schematic structural diagram of a signal modulation system according to an embodiment of the present application;
fig. 2 is a schematic flowchart of a method for suppressing a chirp signal carrier of a radar system according to an embodiment of the present disclosure;
Fig. 3 is a schematic structural diagram of a signal modulation system according to another embodiment of the present application.
Detailed Description
So that the manner in which the features and aspects of the embodiments of the present invention can be understood in detail, a more particular description of the embodiments of the invention, briefly summarized above, may be had by reference to the embodiments, some of which are illustrated in the appended drawings.
in the description of the embodiments of the present application, it should be noted that, unless otherwise specified and limited, the term "connected" should be interpreted broadly, for example, as an electrical connection, a communication between two elements, a direct connection, or an indirect connection via an intermediate, and the specific meaning of the terms may be understood by those skilled in the art according to specific situations.
it should be noted that the terms "first \ second \ third" referred to in the embodiments of the present application are only used for distinguishing similar objects, and do not represent a specific ordering for the objects, and it should be understood that "first \ second \ third" may exchange a specific order or sequence order if allowed. It should be understood that "first \ second \ third" distinct objects may be interchanged under appropriate circumstances such that the embodiments of the application described herein may be implemented in an order other than those illustrated or described herein.
fig. 1 is a schematic structural diagram of a signal modulation system according to an embodiment of the present application, and as shown in fig. 1, the signal modulation system according to the embodiment of the present application includes: a digital baseband circuit 101, a first low-pass filter 102, a first mixer 103, a second low-pass filter 104, a second mixer 105, a local oscillation source 106, a 90 ° power divider 107, and a 0 ° power combiner 108; wherein the digital baseband circuit 101 has an in-phase I-path output end and a quadrature Q-path output end, the digital baseband circuit 101 is connected to the input end of the first low-pass filter 102 through the I-path output end, the digital baseband circuit 101 is connected to the input end of the second low-pass filter 104 through the Q-path output end, the output end of the first low-pass filter 102 is connected to the intermediate frequency IF end of the first mixer 103, the radio frequency RF end of the first mixer 103 is connected to the first input end of the 0 ° power combiner 108, the output end of the second low-pass filter is connected to the IF end of the second mixer, the RF end of the second mixer is connected to the second input end of the 0 ° power combiner, the local oscillator is connected to the input end of the 90 ° power divider, and the first output end of the 90 ° power divider is connected to the LO end of the first mixer, a second output end of the 90 ° power divider is connected to an LO end of the second mixer;
an embodiment of the present application provides a method for suppressing a carrier of a chirp signal of a radar system, which is applied to the signal modulation system, and fig. 2 is a schematic flow diagram of the method for suppressing the carrier of the chirp signal of the radar system according to the embodiment of the present application, and as shown in fig. 2, the method includes:
step 201, applying a first dc bias voltage to the I-band chirp signal sent out by the I-band output terminal.
Fig. 3 is a schematic structural diagram of a signal modulation system according to another embodiment of the present application, and in some embodiments, as shown in fig. 3, the applying a first dc bias voltage to an I-baseband chirp signal sent from the I-channel output terminal includes:
The positive pole of the first direct current voltage is connected with the I-path output end through a first resistor R1, and the negative pole of the first direct current voltage is connected with the I-path output end through a second resistor R2; the positive electrode potential of the first direct-current voltage is + V1, the negative electrode potential of the first direct-current voltage is-V1, and V1 is greater than 0;
The expression of the first dc bias voltage V α is:
The expression of the I-baseband linear frequency modulation signal fI is fI ═ cos (pi kt2), the expression of the local oscillation frequency f η output by the local oscillation source is f η ═ sin (ω t), wherein k is the frequency modulation slope, t is time, and ω is angular frequency;
The expression of the signal fIRF output by the RF terminal of the first mixer after passing through the first mixer is:
f=[cos(πkt)+V]·sin(ωt);
The LO end of the first mixer leaks a signal to the RF end of the first mixer, wherein the signal is p times of f eta, and p is greater than 0;
then, the expression of the carrier f σ output by the RF end of the first mixer is:
And 202, applying a second direct current bias voltage to the Q-path baseband linear frequency modulation signal sent by the Q-path output end.
In some embodiments, as shown in fig. 3, the applying a second dc bias voltage to the Q-branch chirp signal sent from the Q-branch output includes:
the positive pole of the second direct current voltage is connected with the Q-path output end through a third resistor R3, and the negative pole of the second direct current voltage is connected with the Q-path output end through a fourth resistor R4; the positive electrode potential of the second direct-current voltage is + V2, the negative electrode potential of the second direct-current voltage is-V2, and V2 is greater than 0;
the expression of the second dc bias voltage V β is:
The expression of the Q baseband linear frequency modulation signal fQ is fQ ═ sin (pi kt 2);
the expression of the signal fQRF outputted by the RF end of the second mixer after fQ passes through the second mixer is:
f=[sin(πkt)+V]·sin(ωt);
The LO end of the second mixer leaks a signal of the RF end of the second mixer to be q times of f eta, and q is larger than 0;
then, the expression of the carrier f σ output by the RF end of the second mixer is:
Step 203, adjusting the voltage value of the first dc bias voltage within a first preset voltage interval, and detecting a first carrier power variation curve of the carrier power output by the output end of the 0 ° power combiner corresponding to the variation of the voltage value of the first dc bias voltage within the first preset voltage interval.
In some embodiments, the adjusting the voltage value of the first dc bias voltage within the first preset voltage interval includes:
adjusting V1 and/or R1 and/or R2 so that V α varies within the first preset voltage interval.
and 204, taking the voltage value of the first direct current bias voltage corresponding to the lowest carrier power value on the first carrier power variation curve as a first voltage value.
step 205, adjusting the first dc bias voltage to the first voltage value.
Step 206, adjusting the voltage value of the second dc bias voltage within a second preset voltage interval, and detecting a second carrier power variation curve of the carrier power output by the output end of the 0 ° power combiner corresponding to the variation of the voltage value of the second dc bias voltage within the second preset voltage interval.
In some embodiments, the adjusting the voltage value of the second dc bias voltage within the second preset voltage interval includes:
adjusting V2 and/or R3 and/or R4 so that V β varies within the second preset voltage interval.
and step 207, taking the voltage value of the second direct current bias voltage corresponding to the lowest carrier power value on the second carrier power variation curve as a second voltage value.
step 208, adjusting the second dc bias voltage to the second voltage value.
in some embodiments, the above method further comprises:
And adjusting the voltage value of the first direct current bias voltage in the first preset voltage interval, and detecting a third carrier power change curve of the carrier power output by the output end of the 0-degree power combiner, which corresponds to the change of the voltage value of the first direct current bias voltage in the first preset voltage interval.
and taking the voltage value of the first direct current bias voltage corresponding to the lowest carrier power value on the third carrier power variation curve as a third voltage value.
and adjusting the first direct current bias voltage to the third voltage value.
In some embodiments, a specific application of the method for suppressing the chirp carrier of the radar system is provided, wherein R1 is replaced by a first sliding rheostat, so that V α varies within the first preset voltage interval; and replacing R3 with a second sliding rheostat so that V beta is changed within the second preset voltage interval. In particular, the method is applied to a signal modulation system as shown in fig. 3, and comprises the following steps:
Step S1: before the device is powered on to operate, R1 is replaced by a first sliding rheostat, and R3 is replaced by a second sliding rheostat.
step S2: the equipment is powered on to operate, and the quadrature-modulated linear frequency modulation signals are output to a spectrum analyzer to observe the power of the carrier waves.
Step S3: changing the resistance value of the first slide rheostat to increase the resistance value, if the carrier power in the spectrum analyzer is reduced, continuing to increase the resistance value of the first slide rheostat until the carrier power is minimum, and if the carrier power in the spectrum analyzer is increased, reducing the resistance value of the first slide rheostat until the carrier power is minimum.
Step S4: in step S3, the resistance of the second sliding rheostat is changed to increase the resistance, and if the carrier power in the spectrum analyzer is further decreased, the resistance of the second sliding rheostat is continuously increased until the carrier power is minimum, and if the carrier power in the spectrum analyzer is increased, the resistance of the second sliding rheostat is decreased until the carrier power is minimum.
step S5: the first slide rheostat and the second slide rheostat were removed, the resistance values of the two were measured using a multimeter, and fixed resistors having the same resistance values as the slide rheostat were selected and replaced with R1 and R3.
step S6: the equipment is powered on to operate, and the power difference value of the carrier relative to the main signal is read in the spectrum analyzer, so that the carrier suppression degree can be quantitatively measured.
The application has at least the following beneficial effects:
The carrier suppression degree effect is optimal. The carrier suppression of the digital baseband circuit and the carrier suppression of the analog quadrature modulation circuit are unified to compensate each other, and finally the maximum carrier suppression degree of the output linear frequency modulation signal is achieved;
the optimization method is simple and easy to implement. Under the conditions of circuit design and device selection, the optimal carrier suppression degree can be achieved by only replacing 2 resistors, and other key devices such as a mixer, a digital-to-analog converter or an operational amplifier and the like do not need to be replaced;
And the harsh reliability requirement in the production debugging process of the satellite-borne radar is met. In the technical scheme steps of the invention, 2 times of welding operation are carried out on 4 bonding pads in total, the 1 st time is to place the slide rheostat in the circuit, the 2 nd time is to place the fixed resistor with the same resistance value as the slide rheostat in the circuit, the welding times of the same bonding pad meet the aerospace standard, the debugging quantity is small, and the reliability is high.
the features disclosed in the embodiments presented in the present application may be combined arbitrarily, without conflict, to arrive at new method embodiments or apparatus embodiments.
The above description is only for the specific embodiments of the present application, but the scope of the present application is not limited thereto, and any person skilled in the art can easily conceive of the changes or substitutions within the technical scope of the present disclosure, and all the changes or substitutions should be covered by the scope of the present application. Therefore, the protection scope of the present application shall be subject to the protection scope of the claims.

Claims (5)

1. A method for suppressing a chirp signal carrier of a radar system is applied to a signal modulation system, and the signal modulation system is characterized by comprising the following steps: the system comprises a digital baseband circuit, a first low-pass filter, a first frequency mixer, a second low-pass filter, a second frequency mixer, a local oscillation source, a 90-degree power divider and a 0-degree power combiner; wherein the digital baseband circuit has an in-phase I-path output end and a quadrature Q-path output end, the digital baseband circuit is connected to the input end of the first low-pass filter through the I-path output end, the digital baseband circuit is connected to the input end of the second low-pass filter through the Q-path output end, the output end of the first low-pass filter is connected to the intermediate frequency IF end of the first mixer, the radio frequency RF end of the first mixer is connected to the first input end of the 0 ° power combiner, the output end of the second low-pass filter is connected to the IF end of the second mixer, the RF end of the second mixer is connected to the second input end of the 0 ° power combiner, the local oscillator source is connected to the input end of the 90 ° power divider, and the first output end of the 90 ° power divider is connected to the carrier LO end of the first mixer, a second output end of the 90 ° power divider is connected to an LO end of the second mixer;
the method comprises the following steps:
applying a first direct current bias voltage to the I-path linear frequency modulation signal sent by the I-path output end;
Applying a second direct current bias voltage to the Q-path baseband linear frequency modulation signal sent by the Q-path output end;
adjusting the voltage value of the first direct current bias voltage in a first preset voltage interval, and detecting a first carrier power change curve of carrier power output by the output end of the 0-degree power combiner, which corresponds to the change of the voltage value of the first direct current bias voltage in the first preset voltage interval;
taking the voltage value of the first direct current bias voltage corresponding to the lowest value of the carrier power on the first carrier power variation curve as a first voltage value;
Adjusting the first DC bias voltage to the first voltage value;
adjusting the voltage value of the second direct current bias voltage within a second preset voltage interval, and detecting a second carrier power change curve of the carrier power output by the output end of the 0-degree power combiner, which corresponds to the change of the voltage value of the second direct current bias voltage within the second preset voltage interval;
taking the voltage value of the second direct current bias voltage corresponding to the lowest carrier power value on the second carrier power variation curve as a second voltage value;
and adjusting the second direct current bias voltage to the second voltage value.
2. the method of claim 1, wherein said applying a first dc bias voltage to the I-baseband chirp signal from the I-channel output comprises:
The positive pole of the first direct current voltage is connected with the I-path output end through a first resistor R1, and the negative pole of the first direct current voltage is connected with the I-path output end through a second resistor R2; the positive electrode potential of the first direct-current voltage is + V1, the negative electrode potential of the first direct-current voltage is-V1, and V1 is greater than 0;
the expression of the first dc bias voltage V α is:
the expression of the I-baseband linear frequency modulation signal fI is fI ═ cos (pi kt2), the expression of the local oscillation frequency f η output by the local oscillation source is f η ═ sin (ω t), wherein k is the frequency modulation slope, t is time, and ω is angular frequency;
the expression of the signal fIRF output by the RF terminal of the first mixer after passing through the first mixer is:
f=[cos(πkt)+V]·sin(ωt);
the LO end of the first mixer leaks a signal to the RF end of the first mixer, wherein the signal is p times of f eta, and p is greater than 0;
then, the expression of the carrier f σ output by the RF end of the first mixer is:
3. The method of claim 2, wherein applying the second dc bias voltage to the Q-baseband chirp signal from the Q-channel output comprises:
the positive pole of the second direct current voltage is connected with the Q-path output end through a third resistor R3, and the negative pole of the second direct current voltage is connected with the Q-path output end through a fourth resistor R4; the positive electrode potential of the second direct-current voltage is + V2, the negative electrode potential of the second direct-current voltage is-V2, and V2 is greater than 0;
The expression of the second dc bias voltage V β is:
the expression of the Q baseband linear frequency modulation signal fQ is fQ ═ sin (pi kt 2);
the expression of the signal fQRF outputted by the RF end of the second mixer after fQ passes through the second mixer is:
f=[sin(πkt)+V]·sin(ωt);
the LO end of the second mixer leaks a signal of the RF end of the second mixer to be q times of f eta, and q is larger than 0;
Then, the expression of the carrier f σ output by the RF end of the second mixer is:
4. the method of claim 3, wherein the adjusting the voltage value of the first DC bias voltage within a first predetermined voltage interval comprises:
Adjusting V1 and/or R1 and/or R2 so that V α varies within the first preset voltage interval.
5. the method of claim 4, wherein the adjusting the voltage value of the second DC bias voltage within a second predetermined voltage interval comprises:
adjusting V2 and/or R3 and/or R4 so that V β varies within the second preset voltage interval.
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费霞: "X波段宽带线性调频信号的设计与实现", 《中国优秀硕士学位论文全文数据库信息科技辑》 *

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